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FEATURES High Speed 120 MHz Bandwidth, Gain = -1 230 V/ s Slew Rate 90 ns Settling Time to 0.1% Ideal for Video Applications 0.02% Differential Gain 0.04 Differential Phase Low Noise 1.7 nV/Hz Input Voltage Noise 1.5 pA/Hz Input Current Noise Excellent DC Precision 1 mV max Input Offset Voltage (Over Temp) 0.3 V/ C Input Offset Drift Flexible Operation Specified for 5 V to 15 V Operation 3 V Output Swing into a 150 Load External Compensation for Gains 1 to 20 5 mA Supply Current Available in Tape and Reel in Accordance with EIA-481A Standard PRODUCT DESCRIPTION
High-Speed, Low-Noise Video Op Amp AD829
CONNECTION DIAGRAMS 8-Lead Plastic Mini-DIP (N), Cerdip (Q) and SOIC (R) Packages
OFFSET NULL 1 -IN 2 +IN 3 -VS 4
AD829
8 7 6
OFFSET NULL +VS OUTPUT
TOP VIEW 5 CCOMP (Not to Scale)
20-Lead LCC Pinout
NC OFFSET NULL NC OFFSET NULL NC
3 NC 4 -IN 5 NC 6 +IN 7 NC 8 9 10 11 12 13 NC = NO CONNECT 2 1 20 19 18 NC
AD829
TOP VIEW (Not to Scale)
17 +V 16 NC 15 OUTPUT 14 NC
The AD829 is a low noise (1.7 nV/Hz), high speed op amp with custom compensation that provides the user with gains from 1 to 20 while maintaining a bandwidth greater than 50 MHz. The AD829's 0.04 differential phase and 0.02% differential gain performance at 3.58 MHz and 4.43 MHz, driving reverse-terminated 50 or 75 cables, makes it ideally suited for professional video applications. The AD829 achieves its 230 V/s uncompensated slew rate and 750 MHz gain bandwidth product while requiring only 5 mA of current from the power supplies. The AD829's external compensation pin gives it exceptional versatility. For example, compensation can be selected to optimize the bandwidth for a given load and power supply voltage. As a gain-of-two line driver, the -3 dB bandwidth can be increased to 95 MHz at the expense of 1 dB of peaking. In addition, the AD829's output can also be clamped at its external compensation pin. The AD829 has excellent dc performance. It offers a minimum open-loop gain of 30 V/mV into loads as low as 500 , low input voltage noise of 1.7 nV/Hz, and a low input offset voltage of 1 mV maximum. Common-mode rejection and power supply rejection ratios are both 120 dB. The AD829 is also useful in multichannel, high speed data conversion where its fast (90 ns to 0.1%) settling time is of importance. In such applications, the AD829 serves as an input buffer for 8-to-10-bit A/D converters and as an output I/V converter for high speed D/A converters. REV. E
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
The AD829 provides many of the same advantages that a transimpedance amplifier offers, while operating as a traditional voltage feedback amplifier. A bandwidth greater than 50 MHz can be maintained for a range of gains by changing the external compensation capacitor. The AD829 and the transimpedance amplifier are both unity gain stable and provide similar voltage noise performance (1.7 nV/Hz). However, the current noise of the AD829 (1.5 pA/Hz) is less than 10% of the noise of transimpedance amps. Furthermore, the inputs of the AD829 are symmetrical.
PRODUCT HIGHLIGHTS
1. Input voltage noise of 2 nV/Hz, current noise of 1.5 pA/ Hz and 50 MHz bandwidth, for gains of 1 to 20, make the AD829 an ideal preamp. 2. Differential phase error of 0.04 and a 0.02% differential gain error, at the 3.58 MHz NTSC and 4.43 MHz PAL and SECAM color subcarrier frequencies, make it an outstanding video performer for driving reverse-terminated 50 and 75 cables to 1 V (at their terminated end). 3. The AD829 can drive heavy capacitive loads. 4. Performance is fully specified for operation from 5 V to 15 V supplies. 5. Available in plastic, cerdip, and small outline packages. Chips and MIL-STD-883B parts are also available.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 (c) Analog Devices, Inc., 2000
NC CCOMP
NC
NC
-V
AD829-SPECIFICATIONS (@ T = +25 C and V =
A S
15 V dc, unless otherwise noted)
Min AD829J/AR Typ Max 0.2 0.3 3.3 50 0.5 30 20 65 40 30 20 7 8.2/9.5 500 500 1 1 Min AD829AQ/S Typ Max 0.1 0.3 3.3 50 0.5 65 40 50 20 100 85 600 750 25 3.6 150 230 65 90 60 0.02 0.04 100 100 96 98 94 2 120 120 120 1.7 1.5 +4.3 -3.8 +14.3 -13.8 3.0 2.5 12 10 3.6 3.0 1.4 13.3 12.2 32 13 5 1.5 2 2 7 9.5 500 500 0.5 0.5 Units mV mV V/C A A nA nA nA/C V/mV V/mV V/mV V/mV V/mV V/mV MHz MHz MHz MHz V/s V/s ns ns Degrees % Degrees dB dB dB dB dB nV/Hz pA/Hz V V V V V V V V V mA k pF pF m
Model INPUT OFFSET VOLTAGE
Conditions TMIN to TMAX
VS 5 V, 15 V 5 V, 15 V 5 V, 15 V
Offset Voltage Drift INPUT BIAS CURRENT TMIN to TMAX INPUT OFFSET CURRENT TMIN to TMAX Offset Current Drift OPEN-LOOP GAIN VO = 2.5 V RLOAD = 500 TMIN to TMAX RLOAD = 150 VOUT = 10 V RLOAD = 1 k TMIN to TMAX RLOAD = 500
5 V, 15 V 5 V, 15 V 5 V
15 V 50 20 100 85 5 V 15 V 600 750 25 3.6 150 230 65 90 60 15 V 0.02 15 V 0.04 5 V 15 V 100 100 96 98 94 15 V 15 V 5 V 15 V 120 120 120 1.7 1.5 +4.3 -3.8 +14.3 -13.8 3.0 2.5 12 10 3.6 3.0 1.4 13.3 12.2 32 13 5 1.5
DYNAMIC PERFORMANCE Gain Bandwidth Product Full Power Bandwidth1, 2 VO = 2 V p-p RLOAD = 500 VO = 20 V p-p RLOAD = 1 k RLOAD = 500 RLOAD = 1 k AV = -19 -2.5 V to +2.5 V 10 V Step CLOAD = 10 pF RLOAD = 1 k
3
5 V 15 V 5 V 15 V 5 V 15 V 15 V
Slew Rate2 Settling Time to 0.1%
Phase Margin2 DIFFERENTIAL GAIN ERROR
RLOAD = 100 CCOMP = 30 pF
3
DIFFERENTIAL PHASE ERROR COMMON-MODE REJECTION
RLOAD = 100 CCOMP = 30 pF VCM = 2.5 V VCM = 12 V TMIN to TMAX VS = 4.5 V to 18 V TMIN to TMAX f = 1 kHz f = 1 kHz
POWER SUPPLY REJECTION INPUT VOLTAGE NOISE INPUT CURRENT NOISE INPUT COMMON-MODE VOLTAGE RANGE
OUTPUT VOLTAGE SWING
RLOAD = 500 RLOAD = 150 RLOAD = 50 RLOAD = 1 k RLOAD = 500
Short Circuit Current INPUT CHARACTERISTICS Input Resistance (Differential) Input Capacitance (Differential)4 Input Capacitance (Common Mode) CLOSED-LOOP OUTPUT RESISTANCE AV = +1, f = 1 kHz
5 V 5 V 5 V 15 V 15 V 5 V, 15 V
2
-2-
REV. E
AD829
Model POWER SUPPLY Operating Range Quiescent Current TMIN to TMAX TMIN to TMAX TRANSISTOR COUNT Number of Transistors 46 Conditions VS Min 4.5 5 5.3 AD829J/AR Typ Max Min AD829AQ/S Typ Max 18 6.5 8.2/8.7 6.8 8.5/9.0 Units V mA mA mA mA 18 4.5 6.5 8.0 6.8 8.3/8.5
5 V 15 V
5 5.3 46
NOTES 1 Full Power Bandwidth = Slew Rate/2 VPEAK. 2 Tested at Gain = +20, C COMP = 0 pF. 3 3.58 MHz (NTSC) and 4.43 MHz (PAL & SECAM). 4 Differential input capacitance consists of 1.5 pF package capacitance plus 3.5 pF from the input differential pair. Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS 1
METALIZATION PHOTO
Contact factory for latest dimensions. Dimensions shown in inches and (mm).
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipations2 Plastic (N) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts Small Outline (R) . . . . . . . . . . . . . . . . . . . . . . . . . 0.9 Watts Cerdip (Q) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts LCC (E) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.8 Watts Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VS Differential Input Voltage3 . . . . . . . . . . . . . . . . . . . . 6 Volts Output Short Circuit Duration . . . . . . . . . . . . . . . . Indefinite Storage Temperature Range (Q, E) . . . . . . . -65C to +150C Storage Temperature Range (N, R) . . . . . . . -65C to +125C Operating Temperature Range AD829J . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to +70C AD829A . . . . . . . . . . . . . . . . . . . . . . . . . . . -40C to +85C AD829S . . . . . . . . . . . . . . . . . . . . . . . . . . -55C to +125C Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300C
NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Maximum internal power dissipation is specified so that T J does not exceed +175C at an ambient temperature of +25C. Thermal characteristics: 8-lead plastic package: JA = 100C/watt (derate at 8.7 mW/C) 8-lead cerdip package: JA = 110C/watt (derate at 8.7 mW/C) 20-lead LCC package: JA = 150C/watt 8-lead small outline package: JA = 155C/watt (derate at 6 mW/C). 3 If the differential voltage exceeds 6 volts, external series protection resistors should be added to limit the input current.
ESD SUSCEPTIBILITY
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 volts, which readily accumulate on the human body and on test equipment, can discharge without detection. Although the AD829 features proprietary ESD protection circuitry, permanent damage may still occur on these devices if they are subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid any performance degradation or loss of functionality.
ORDERING GUIDE
Model AD829JN AD829AR AD829JR AD829AR-REEL7 AD829AR-REEL AD829JR-REEL7 AD829JR-REEL AD829AQ AD829SQ AD829SQ/883B 5962-9312901MPA AD829SE/883B 5962-9312901M2A AD829JCHIPS AD829SCHIPS REV. E
Temperature Range 0C to +70C -40C to +85C 0C to +70C -40C to +85C -40C to +85C 0C to +70C 0C to +70C -40C to +85C -55C to +125C -55C to +125C -55C to +125C -55C to +125C -55C to +125C 0C to +70C -55C to +125C -3-
Package Description 8-Lead Plastic Mini-DIP 8-Lead Plastic SOIC 8-Lead Plastic SOIC Tape and Reel 7" Tape and Reel 13" Tape and Reel 7" Tape and Reel 13" 8-Lead Cerdip 8-Lead Cerdip 8-Lead Cerdip 8-Lead Cerdip 20-Lead LCC 20-Lead LCC Die Die
Package Option* N-8 SO-8 SO-8
Q-8 Q-8 Q-8 Q-8 E-20A E-20A
*E = Leadless Chip Carrier (Ceramic); N = Plastic DIP; Q = Cerdip; SO = Small Outline IC (SOIC).
AD829-Typical Performance Characteristics
20
INPUT COMMON-MODE RANGE - Volts
20
OUTPUT VOLTAGE SWING - Volts p-p
30 15 VOLT SUPPLIES MAGNITUDE OF THE OUTPUT VOLTAGE - Volts 25
15 +VOUT 10 -VOUT 5
15 +VOUT
20
10 -VOUT
15
10 5 0 10 5 VOLT SUPPLIES
5 RLOAD = 1k 0
0
0
5 10 SUPPLY VOLTAGE -
15 Volts
20
0
5 10 SUPPLY VOLTAGE -
15 Volts
20
1k 100 LOAD RESISTANCE -
10k
Figure 1. Input Common-Mode Range vs. Supply Voltage
Figure 2. Output Voltage Swing vs. Supply Voltage
Figure 3. Output Voltage Swing vs. Resistive Load
6.0
-5
CLOSED - LOOP OUTPUT IMPEDANCE -
100
QUIESCENT CURRENT - mA
A
INPUT BIAS CURRENT -
5.5
10
-4
AV = +20 CCOMP = 0pF
1
5.0
VS = -3
5V,
15V
0.1 AV = +1 CCOMP = 68pF 0.01
4.5
4.0
0
5 10 SUPPLY VOLTAGE -
15 Volts
20
-2 - 60 - 40 - 20
0 20 40 60 80 100 120 140 TEMPERATURE - C
0.001 1k
10k
100k 1M 10M FREQUENCY - Hz
100M
Figure 4. Quiescent Current vs. Supply Voltage
Figure 5. Input Bias Current vs. Temperature
Figure 6. Closed-Loop Output Impedance vs. Frequency
7
40
65 NEGATIVE CURRENT LIMIT VS = 15V AV = +20 CCOMP = 0pF 60
SHORT CIRCUIT CURRENT LIMIT - mA
QUIESCENT CURRENT - mA
6 VS = 15V
30
POSITIVE CURRENT LIMIT
-3 dB BANDWIDTH - MHz
35
5 VS = 4 5V
55
25
VS =
5V
50
20
3 - 60 - 40 - 20
0 20 40 60 80 100 120 140 TEMPERATURE - C
15 - 60 - 40 - 20 0 20 40 60 80 100 120 140 AMBIENT TEMPERATURE - C
45 - 60 - 40 - 20
0 20 40 60 80 100 120 140 TEMPERATURE - C
Figure 7. Quiescent Current vs. Temperature
Figure 8. Short Circuit Current Limit vs. Temperature
Figure 9. -3 dB Bandwidth vs. Temperature
-4-
REV. E
AD829
120 PHASE 100
OPEN-LOOP GAIN - dB
+100
105 100
OPEN-LOOP GAIN - dB
120 +SUPPLY
+80
PHASE - Degrees
80
PSRR - dB
GAIN 15V Supplies 1k Load GAIN 5V Supplies 500 Load CCOMP = 0pF
VS = 95
15V
100
+60
80
- SUPPLY
60
+40
90
VS =
5V
60
40
+20
85 40 CCOMP = 0pF
20 0 100
0 -20 100M
80
1k
10k 100k 1M FREQUENCY - Hz
10M
75 10
1k 100 LOAD RESISTANCE -
10k
20 1k
10k
100k 1M 10M FREQUENCY - Hz
100M
Figure 10. Open-Loop Gain & Phase Margin vs. Frequency
Figure 11. Open-Loop Gain vs. Resistive Load
Figure 12. Power Supply Rejection Ratio (PSRR) vs. Frequency
120
OUTPUT VOLTAGE - Volts p-p
30 25 VS = 15V RL = 1k AV = +20 CCOMP = 0pF VS = 5V RL = 500 AV = +20 CCOMP = 0pF
V OUTPUT SWING FROM 0 TO
10 8
100
6 4 2 1% 0 1% -2 -4 -6 -8 0.1% 0.1% ERROR AV = -19 CCOMP = 0pF
20
CMRR - dB
80
15
60 CCOMP = 0pF 40
10
5 0
20 1k
-10
10k
100k 1M FREQUENCY - Hz
10M
100M
1
10 INPUT FREQUENCY - MHz
100
0
20
40 60 80 100 120 140 160 SETTLING TIME - ns
Figure 13. Common-Mode Rejection Ratio vs. Frequency
Figure 14. Large Signal Frequency Response
Figure 15. Output Swing & Error vs. Settling Time
-70 -75 -80
THD - dB
-20
INPUT VOLTAGE NOISE - nV/ Hz
5
VIN = 2.24V RMS AV = -1 RL = 250 CLOAD = 0 CCOMP = 30pF 3rd HARMONIC
VIN = 3V RMS AV = -1 CCOMP = 30pF CLOAD = 100pF RL = 500
THD - dB
-30
4
-85 -90 -95
-40
3
-50 2nd HARMONIC
2
-100 RL = 2k -105 -110 100 300 1k 3k 10k FREQUENCY - Hz 30k 100k
-60
1
-70 0
500k 1M 1.5M FREQUENCY - Hz
2M
0 10
100
1k 10k 100k FREQUENCY - Hz
1M
10M
Figure 16. Total Harmonic Distortion (THD) vs. Frequency
Figure 17. 2nd & 3rd Harmonic Distortion vs. Frequency
Figure 18. Input Voltage Noise Spectral Density
REV. E
-5-
AD829-Typical Performance Characteristics
400 DIFFERENTIAL PHASE - Degrees AV = +20 SLEW RATE 10 - 90% RISE
0.03 0.02
DIFFERENTIAL GAIN - Percent
350
CCOMP (EXTERNAL)
+VS
0.1 F
SLEW RATE - Volts / s
300
DIFF GAIN
0.01
250 200
VS =
15V
FALL RISE FALL
AD829
0.1 F 20k
0.043 0.05 DIFF PHASE 0.04 0.03
150 VS = 100 - 60 - 40 - 20 5V
0 20 40 60 80 100 120 140 TEMPERATURE - C
5
10 SUPPLY VOLTAGE - Volts
15
OFFSET NULL ADJUST
-VS
Figure 19. Slew Rate vs. Temperature
Figure 20. Differential Gain & Phase vs. Supply
Figure 21. Offset Null and External Shunt Compensation Connections
0.1 F
+15V
CCOMP 15pF
50 CABLE HP8130A 5ns RISE TIME 50 50 50 CABLE
AD829
TEKTRONIX TYPE 7A24 PREAMP 50
0.1 F
5pF
300
-15V 300
Figure 22a. Follower Connection. Gain = +2
Figure 22b. Gain-of-2 Follower Large Signal Pulse Response
Figure 22c. Gain-of-2 Follower Small Signal Pulse Response
-6-
REV. E
AD829
+15V 0.1 F 50 CABLE HP8130A 5ns RISE TIME
45 5
100
FET PROBE
AD829
TEKTRONIX TYPE 7A24 PREAMP 1pF 2k
0.1 F
-15V CCOMP = 0pF 105
Figure 23a. Follower Connection. Gain = +20
Figure 23b. Gain-of-20 Follower Large Signal Pulse Response
5pF 300 +15V 0.1 F 50 CABLE
Figure 23c. Gain-of-20 Follower Small Signal Pulse Response
HP8130A 5ns RISE TIME
300
50
AD829
CCOMP 15pF
50
50 CABLE
TEKTRONIX TYPE 7A24 PREAMP 50
0.1 F -15V
Figure 24a. Unity Gain Inverter Connection
Figure 24b. Unity Gain Inverter Large Signal Pulse Response
Figure 24c. Unity Gain Inverter Small Signal Pulse Response
REV. E
-7-
AD829
THEORY OF OPERATION
+VS
The AD829 is fabricated on Analog Devices' proprietary complementary bipolar (CB) process which provides PNP and NPN transistors with similar fTs of 600 MHz. As shown in Figure 25, the AD829 input stage consists of an NPN differential pair in which each transistor operates at 600 A collector current. This gives the input devices a high transconductance and hence gives the AD829 a low noise figure of 2 nV/Hz @ 1 kHz. The input stage drives a folded cascode which consists of a fast pair of PNP transistors. These PNPs then drive a current mirror which provides a differential-input to single-ended-output conversion. The high speed PNPs are also used in the currentamplifying output stage which provides high current gain of 40,000. Even under conditions of heavy loading, the high fTs of the NPN & PNPs, produced using the CB process, permit cascading two stages of emitter followers while still maintaining 60 of phase margin at closed-loop bandwidths greater than 50 MHz. Two stages of complementary emitter followers also effectively buffer the high impedance compensation node (at the CCOMP pin) from the output so that the AD829 can maintain a high dc open-loop gain, even into low load impedances: 92 dB into a 150 load, 100 dB into a 1 k load. Laser trimming and PTAT biasing assure low offset voltage and low offset voltage drift enabling the user to eliminate ac coupling in many applications. For added flexibility, the AD829 provides access to the internal frequency compensation node. This allows the user to customize frequency response characteristics for a particular application. Unity gain stability requires a compensation capacitance of 68 pF (Pin 5 to ground) which will yield a small signal bandwidth of 66 MHz and slew rate of 16 V/s. The slew rate and gain bandwidth product will vary inversely with compensation capacitance. Table I and the graph of Figure 28 show the optimum compensation capacitance and the resulting slew rate for a desired noise gain. For gains between 1 and 20, CCOMP can be chosen to keep the small signal bandwidth relatively constant. The minimum gain which will still provide stability also depends on the value of external compensation capacitance. An RC network in the output stage (Figure 25) completely removes the effect of capacitive loading when the amplifier is compensated for closed-loop gains of 10 or higher. At low frequencies, and with low capacitive loads, the gain from the compensation node to the output is very close to unity. In this case, C is bootstrapped and does not contribute to the compensation capacitance of the device. As the capacitive load is increased, a pole is formed with the output impedance of the output stage- this reduces the gain, and subsequently, C is incompletely bootstrapped. Therefore, some fraction of C contributes to the compensation capacitance, and the unity gain bandwidth falls. As the load capacitance is further increased, the bandwidth continues to fall, and the amplifier remains stable.
Externally Compensating the AD829
+IN -IN
15 OUTPUT C 12.5pF R 500 15
1.2mA
-VS OFFSET NULL CCOMP
Figure 25. AD829 Simplified Schematic
Shunt Compensation
Figures 26 and 27 show that the first method, shunt compensation, has an external compensation capacitor, CCOMP, connected between the compensation pin and ground. This external capacitor is tied in parallel with approximately 3 pF of internal capacitance at the compensation node. In addition, a small capacitance, CLEAD, in parallel with resistor R2, compensates for the capacitance at the amplifier's inverting input.
R2 CLEAD +VS 50 COAX CABLE VIN 50 0.1 F R1
AD829
CCOMP 0.1 F 1k
VOUT
-VS
Figure 26. Inverting Amplifier Connection Using External Shunt Compensation
+VS 0.1 F 50 CABLE VIN 50
AD829
R2 CCOMP -VS 0.1 F 1k
VOUT
CLEAD
The AD829 is stable with no external compensation for noise gains greater than 20. For lower gains, there are two methods of frequency compensating the amplifier to achieve closed-loop stability; these are the shunt and current feedback compensation methods.
R1
Figure 27. Noninverting Amplifier Connection Using External Shunt Compensation
-8-
REV. E
AD829
Table I. Component Selection for Shunt Compensation
Follower Gain 1 2 5 10 20 25 100
Inverter Gain -1 -4 -9 -19 -24 -99
R1 Open 1k 511 226 105 105 20
R2 100 1k 2.0k 2.05k 2k 2.49 2k
CL pF 0 5 1 0 0 0 0
CCOMP pF 68 25 7 3 0 0 0 then:
Slew Rate V/ s 16 38 90 130 230 230 230
-3 dB Small Signal Bandwidth - MHz 66 71 76 65 55 39 7.5
Table I gives recommended CCOMP and CLEAD values along with the corresponding slew rates and bandwidth. The capacitor values given were selected to provide a small signal frequency response with less than 1 dB of peaking and less than 10% overshoot. For this table, supply voltages of 15 volts should be used. Figure 28 is a graphical extension of the table which shows the slew rate/gain trade-off for lower closed-loop gains, when using the shunt compensation scheme.
100 1k
Slew Rate kT =4 q fT
This shows that the slew rate will be only 0.314 V/s for every MHz of bandwidth. The only way to increase slew rate is to increase the fT and that is difficult, due to process limitations. Unfortunately, an amplifier with a bandwidth of 10 MHz can only slew at 3.1 V/s, which is barely enough to provide a full power bandwidth of 50 kHz. The AD829 is especially suited to a new form of compensation which allows for the enhancement of both the full power bandwidth and slew rate of the amplifier. The voltage gain from the inverting input pin to the compensation pin is large; therefore, if a capacitance is inserted between these pins, the amplifier's bandwidth becomes a function of its feedback resistor and this capacitance. The slew rate of the amplifier is now a function of its internal bias (2I) and this compensation capacitance. Since the closed-loop bandwidth is a function of RF and CCOMP (Figure 29), it is independent of the amplifier closed-loop gain, as shown in Figure 31. To preserve stability, the time constant of RF and CCOMP needs to provide a bandwidth of less than 65 MHz. For example, with CCOMP = 15 pF and RF = 1 k, the small signal bandwidth of the AD829 is 10 MHz, while Figure 30 shows that the slew rate is in excess of 60 V/s. As can be seen in Figure 31, the closed-loop bandwidth is constant for gains of -1 to -4, a property of current feedback amplifiers.
RF
10
100
VS =
15V
1 1 10 NOISE GAIN
10 100
Figure 28. Value of CCOMP & Slew Rate vs. Noise Gain
Current Feedback Compensation
Bipolar nondegenerated amplifiers which are single pole and internally compensated have their bandwidths defined as:
1 fT = = 2 r e CCOMP
I kT 2 q CCOMP
50 COAX CABLE VIN 50 R1 C1* IN4148
SLEW RATE = V/ s
CCOMP
CCOMP - pF
SLEW RATE
CCOMP 0.1 F +V S
where: fT is the unity gain bandwidth of the amplifier I is the collector current of the input transistor CCOMP is the compensation capacitance re is the inverse of the transconductance of the input transistors kT/q is approximately equal to 26 mV @ 27C. Since both fT and slew rate are functions of the same variables, the dynamic behavior of an amplifier is limited. Since:
AD829
0.1 F RL 1k
VOUT
*RECOMMENDED VALUE OF CCOMP FOR C1 <7pF 7pF 0pF 15pF
-VS CCOMP SHOULD NEVER EXCEED 15pF FOR THIS CONNECTION
Slew Rate =
2I CCOMP
Figure 29. Inverting Amplifier Connection Using Current Feedback Compensation
REV. E
-9-
AD829
Figure 30. Large Signal Pulse Response of Inverting Amplifier Using Current Feedback Compensation. CCOMP = 15 pF, C1 = 15 pF, RF = 1 k, R1 = 1 k
15 12 9 GAIN = -4 -3dB @ 8.2MHz
Figure 32. Large Signal Pulse Response of the Inverting Amplifier Using Current Feedback Compensation. CCOMP = 1 pF, RF = 3 k, R1 = 3 k
CLOSED-LOOP GAIN - dB
GAIN = -2 6 3 GAIN = -1 0 -3dB @ 10.2MHz -3 -6 -9 -12 -15 100k VIN = -30dBM VS = 15V RL = 1k RF = 1k CCOMP = 15pF C1 = 15pF 1M 10M FREQUENCY - Hz 100M -3dB @ 9.6MHz
Figure 31. Closed-Loop Gain vs. Frequency for the Circuit of Figure 29
Figure 33. Small Signal Pulse Response of Inverting Amplifier Using Current Feedback Compensation. CCOMP = 4 pF, RF = 1 k, R1 = 1 k
15 12 9 GAIN = -4 CCOMP = 2pF
CLOSED-LOOP GAIN - dB
Figure 32 is an oscilloscope photo of the pulse response of a unity gain inverter which has been configured to provide a small signal bandwidth of 53 MHz and a subsequent slew rate of 180 V/s; resistor RF = 3 k, capacitor CCOMP = 1 pF. Figure 33 shows the excellent pulse response as a unity gain inverter, this time using component values of: RF = 1 k and CCOMP = 4 pF. Figures 34 and 35 show the closed-loop frequency response of the AD829 for different closed-loop gains and for different supply voltages. If a noninverting amplifier configuration using current feedback compensation is desired, the circuit of Figure 36 is recommended. This circuit doubles the slew rate compared to the shunt compensated noninverting amplifier of Figure 27 at the expense of gain flatness. Nonetheless, this circuit delivers 95 MHz bandwidth with 1 dB flatness into a back terminated cable, with a differential gain error of only 0.01%, and a differential phase error of only 0.015 at 4.43 MHz.
GAIN = -2 6 3 GAIN = -1 0 -3 -6 -9 -12 -15 1M VS = 15V RL = 1k RF = 1k VIN = -30dBM
CCOMP = 3pF
CCOMP = 4pF
10M FREQUENCY - Hz
100M
Figure 34. Closed-Loop Frequency Response for the Inverting Amplifier Using Current Feedback Compensation
-10-
REV. E
AD829
-17 -20 -23 5V
+15V 0.1 F 50 COAX CABLE VIN 50 50 COAX CABLE VOUT 50
OUTPUT LEVEL - dB
-26 15V -29 -32 -35 -38 -41 -44 -47 1M VIN = -20dBM RL = 1k RF = 1k GAIN = -1 CCOMP = 4pF 10M FREQUENCY - Hz 100M
AD829
50
-15V 0.1 F
3pF CCOMP
2k
2k
Figure 36. Noninverting Amplifier Connection Using Current Feedback Compensation
+15V 0.1 F 75 COAX CABLE VOUT 75 0.1 F 30pF CCOMP -15V 300
Figure 35. Closed-Loop Frequency Response vs. Supply for the Inverting Amplifier Using Current Feedback Compensation
A Low Error Video Line Driver
VIN
The buffer circuit shown in Figure 37 will drive a back-terminated 75 video line to standard video levels (1 V p-p) with 0.1 dB gain flatness to 30 MHz with only 0.04 and 0.02% differential phase and gain at the 4.43 MHz PAL color subcarrier frequency. This level of performance, which meets the requirements for high definition video displays and test equipment, is achieved using only 5 mA quiescent current.
A High Gain, Video Bandwidth Three Op Amp In Amp
AD829
75
75
OPTIONAL 2 - 7pF FLATNESS TRIM
300
Figure 38 shows a three op amp instrumentation amplifier circuit which provides a gain of 100 at video bandwidths. At a circuit gain of 100 the small signal bandwidth equals 18 MHz into an FET probe. Small signal bandwidth equals 6.6 MHz with a 50 load. 0.1% settling time is 300 ns.
Figure 37. A Video Line Driver with a Flatness over Frequency Adjustment
3pF +VIN A1 (G = 20) 2-8pF SETTLING TIME AC CMR ADJUST
The input amplifiers operate at a gain of 20, while the output op amp runs at a gain of 5. In this circuit the main bandwidth limitation is the gain/ bandwidth product of the output amplifier. Extra care needs to be taken while breadboarding this circuit, since even a couple of extra picofarads of stray capacitance at the compensation pins of A1 and A2 will degrade circuit bandwidth.
AD829
2k 1pF RG 210 200
1k
AD848
A3 (G = 5) 3pF 970 2k INPUT FREQUENCY 100 Hz 1 MHz 10 MHz +15V 10 F COMM CIRCUIT GAIN = 0.1 F CMRR 64.6dB 44.7dB 23.9dB +VS 1F PIN 7 0.1 F EACH AMPLIFIER PIN 4
1pF 2k
200
AD829
A2 +VIN (G = 20) 3pF
DC CMR ADJUST 50
( 4000 RG
+1 5
(
10 F -15V
0.1 F -VS
1F
0.1 F
Figure 38. A High Gain, Video Bandwidth Three Op Amp In Amp Circuit
REV. E
-11-
AD829
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
Cerdip (Q) Package
0.005 (0.13) MIN 0.055 (1.40) MAX
8 PIN 1 1
5 0.310 (7.87) 0.220 (5.59) 4 0.320 (8.13) 0.290 (7.37) 0.060 (1.52) 0.015 (0.38)
0.405 (10.29) MAX 0.200 (5.08) MAX 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.100 0.070 (1.78) 0.014 (0.36) (2.54) 0.030 (0.76) BSC
0.150 (3.81) MIN
0.015 (0.38) 0.008 (0.20) 15 0
SEATING PLANE
Plastic Mini-DIP (N) Package
8-Lead SOIC (R) Package
0.1968 (5.00) 0.1890 (4.80)
8 5 4
8 PIN 1 1
5 0.25 (6.35) 0.31 (7.87) 4 0.30 (7.62) REF 0.0350.01 (0.890.25)
0.1574 (4.00) 0.1497 (3.80) PIN 1
1
0.2440 (6.20) 0.2284 (5.80)
0.39 (9.91) MAX
0.1650.01 (4.190.25) 0.125 (3.18) MIN 0.0180.003 (0.460.08) 0.10 (2.54) BSC 0.033 (0.84) NOM
0.0500 (1.27) BSC 0.0098 (0.25) 0.0040 (0.10) SEATING PLANE 0.0688 (1.75) 0.0532 (1.35) 0.0192 (0.49) 0.0138 (0.35) 8 0.0098 (0.25) 0 0.0075 (0.19)
0.0196 (0.50) 0.0099 (0.25)
45
0.180.03 (4.570.76)
0.0110.003 (0.280.08) 15 0
0.0500 (1.27) 0.0160 (0.41)
SEATING PLANE
20-Lead LCC (E-20A) Package
0.200 (5.08) BSC 0.100 (2.54) BSC
3 4 1
0.100 (2.54) 0.064 (1.63) 0.095 (2.41) 0.075 (1.90) 0.011 (0.28) 0.007 (0.18) R TYP 0.075 (1.91) REF
0.075 (1.91) REF
19 18 20
0.015 (0.38) MIN
BOTTOM VIEW
14 13 8 9
0.050 (1.27) BSC
0.088 (2.24) 0.054 (1.37)
0.055 (1.40) 0.045 (1.14)
45 TYP 0.150 (3.81) BSC
All brand or product names mentioned are trademarks or registered trademarks of their respective holders.
-12-
REV. E
PRINTED IN U.S.A.
0.358 (9.09) 0.358 (9.09) 0.342 (8.69) MAX SQ SQ
0.028 (0.71) 0.022 (0.56)
C1443c-0-5/00 (rev. E) 00880
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